View SP765507_1315148.PDF datasheet online --- IC-ON-LINE

Datasheet File OCR Text:

Wide Input Voltage Range, 8Amp 300kHz, Buck Regulator
FEATURES

SP7655
TM
Power Blox
DFN PACKAGE 7mm x 4m m
SP7655
2.5V to 28V Step Down Achieved Using Dual Input Output Voltage down to 0.8V 8A Output Capability Built in Low RDSON Power Switches (15m typ) Highly Integrated Design, Minimal Components 300 kHz Fixed Frequency Operation UVLO Detects Both VCC and V IN Over Temperature Protection Short Circuit Protection with Auto-Restart Wide BW Amp Allows Type II or III Compensation Programmable Soft Start Fast Transient Response High Efficiency: Greater than 95%Possible Non-synchronous Start-Up into a Pre-Charged Output Lead Free, RoHS Compliant Package: Small 7mm x 4mm DFN U.S. Patent #6,922,041
PGND 1 PGND 2 PGND 3 GND 4 VFB 5 COMP 6 UVIN 7 GND 8 SS 9 VIN 10 VIN 11 VIN 12 VIN 13 Heatsink Pad 3 Connect toVIN Heatsink Pad 2 Connect to GND TOP VIEW Heatsink Pad 1 Connect to Lx
26 LX 25 LX 24 LX 23 LX 2 2 VCC 21 GND 20 GND 19 GND 18 BST 17 NC 16 LX 15 LX 14 LX
The SP7655 is a synchronous step-down switching regulator optimized for high efficiency. The part is designed to be especially attractive for dual supply, 12V or 24V distributed power systems stepped down with 5V used to power the controller. This lower VCCvoltage minimizes power dissipation in the part and is used to drive the top switch. The SP7655 is designed to provide a fully integrated buck regulator solution using a fixed 300kHz frequency, PWM voltage mode architecture. Protection features include UVLO, thermal shutdown and output short circuit protection. The SP7655 is available in the space saving DFN package.
DESCRIPTION
TYPICAL APPLICATION CIRCUIT
U1 SP7655
1 PGND PGND PGND GND VFB COMP UVIN GND SS VIN VIN VIN VIN LX LX LX LX VCC GND GND GND BST NC LX LX LX 26 25 24 23 22 21 20 19 18 17 16 15 14
L1
2.2uH, Irate=8A
CZ2
RZ2 CP1
22pF
2 3
1,000pF 15k ,1%
4 5 6
+5V VCC CVCC
2.2uF
C3
47uF 6.3V
RZ3
7.15k 1%
VOUT 3.30V 0-8A
68.1k ,1%
CZ3
150pF
R1
100pF
CF1
ENABLE
7 8 9
DBST
SD101AWS 6,800pF
RSET
21.5k ,1% (note 2)
CSS
47nF
10 11 12
CBST
VIN 12V
22uF 16V
13
C1
fs=300Khz
22uF 16V
C4
Notes:
1. U1 Bottom side layout should have three contacts isolated from one another: Vin, SWNODE, and GND. 2. RSET=54.48/(Vout-0.8V) KOhms
GND
Rev E: 3/2/07
SP7655 Wide Input Voltage Range 8A, 300kHz, Buck Regulator
(c) 2007 Sipex Corporation
ABSOLUTE MAXIMUM RATINGS
These are stress ratings only and functional operation of the device at these ratings or any other above those indicated in the operation sections of the specifications below is not implied. Exposure to absolute maximum rating conditions for extended periods of time may affect reliability. VCC .................................................................................................. 7V VIN ........................................................................................................................................... 30V ILX ............................................................................................................................................ 10A BST ............................................................................................... 35V LX-BST ............................................................................. -0.3V to 7V LX ....................................................................................... -1V to 30V All other pins .......................................................... -0.3V to VCC+0.3V Storage Temperature .................................................. -65C to 150C Power Dissipation ...................................... Internally Limited via OTP ESD Rating .......................................................................... 2kV HBM Thermal Resistance JC .................................................................................... 5C/W
Unless otherwise specified: -40C < TAMB< 85 C, -40C < Tj< 125C, 4.5V < VCC < 5.5V, 3V< Vin< 28V, BST=LX + 5V, LX = GND = 0.0V, UVIN = 3.0V, CVCC= 1 F, CCOMP= 0.1 F, CSS= 50nF, Typical measured at V CC = 5V. The denotes the specifications which apply over the full temperature range, unless otherwise specified.
ELECTRICAL SPECIFICATIONS
PARAMETER QUIESCENT CURRENT VCC Supply Current (No switching) VCC Supply Current (switching) BST Supply Current (No switching) BST Supply Current (switching) PROTECTION: UVLO VCC UVLO Start Threshold VCC UVLO Hysteresis UVIN Start Threshold UVIN Hysteresis UVIN Input Current ERROR AMPLIFIER REFERENCE Error Amplifier Reference Error Amplifier Reference Over Line and Temperature Error Amplifier Transconductance Error Amplifier Gain COMP Sink Current COMP Source Current VFB Input Bias Current Internal Pole COMP Clamp COMP Clamp Temp. Coefficient
Rev E: 3/2/07
MIN.
TYP.
MAX.
UNITS
CONDITIONS
1.5 6 0.2 5
3 9 0.4 8
mA mA mA mA
VFB =0.9V
VFB =0.9V
4.00 100 2.3 200
4.25 200 2.5 300
4.5 300 2.65 400 1
V mV V mV A UVIN =3.0V
0.792 0.788
0.800 0.800 6 60 150 150 50 4 2.5 -2
0.808 0.812
V V mA/V dB A A
2X Gain Config., Measure VFB; VCC =5 V, T= 25C
No Load VFB =0.9V, COMP=0.9V VFB =0.7V, COMP=2..2V VFB =0.8V
200
nA MHz V mV/C
VFB =0.7V, TA=25C
SP7655 Wide Input Voltage Range 8A, 300kHz, Buck Regulator
(c) 2007 Sipex Corporation
2
Unless otherwise specified: -40C < TAMB< 85 C, -40C< Tj< 125C, 4.5V < VCC < 5.5V, 3V< Vin< 28V, BST=LX + 5V, LX = GND = 0.0V, UVIN = 3.0V, CVCC= 1 F, CCOMP= 0.1 F, CSS= 50nF, Typical measured at V CC = 5V. The denotes the specifications which apply over the full temperature range, unless otherwise specified.
ELECTRICAL SPECIFICATIONS
PARAMETER
MIN.
TYP.
MAX.
UNITS
CONDITIONS
CONTROL LOOP: PWM COMPARATOR, RAMP & LOOP DELAY PATH Ramp Amplitude RAMP Offset RAMP Offset Temp. Coefficient GH Minimum Pulse Width Maximum Controllable Duty Ratio Maximum Duty Ratio Internal Oscillator Ratio TIMERS: SOFTSTART SS Charge Current: SS Discharge Current: PROTECTION: Short Circuit & Thermal Short Circuit Threshold Voltage Hiccup Timeout Number of Allowable Clock Cycles at 100% Duty Cycle Minimum GL Pulse After 20 Cycles Thermal Shutdown Temperature Thermal Recovery Temperature Thermal Hysteresis OUTPUT: POWER STAGE High Side RDSON Synchronous FET RDSON Maximum Output Current
Rev E: 3/2/07
0.92
1.1 1.1 -2 90
1.28
V V mV/C TA = 25C, RAMP COMP until GH starts Switching
18 0
ns % %
Maximum Duty Ratio Measured just before pulsing begins Valid for 20 cycles
92 100 240
97
300
360
kHz
10 1
A mA Fault Present, SS = 0.2V
0.2
0.25 200 20 0.5 145 135 10
0.3
V ms Cycles Cycles C C C
Measured VREF (0.8V) V FB VFB =0.5V
VFB =0.7V VFB =0.7V
15 15 8
m m A

VCC = 5V ; IOUT = 8A TAMB = 25C VCC = 5V ; IOUT = 8A TAMB = 25C
SP7655 Wide Input Voltage Range 8A, 300kHz, Buck Regulator
(c) 2007 Sipex Corporation
3
PIN DESCRIPTION
Pin # 1-3 4,8,19-21 Pin Name PGND GND Description Ground connection for the synchronous rectifier Ground Pin. The control circuitry of the IC and lower power driver are referenced to this pin. Return separately from other ground traces to the (-) terminal of COUT. Feedback Voltage and Short Circuit Detection pin. It is the inverting input of the Error Amplifier and serves as the output voltage feedback point for the Buck Converter. The output voltage is sensed and can be adjusted through an external resistor divider. Whenever V FB drops 0.25V below the positive reference, a short circuit fault is detected and the IC enters hiccup mode. Output of the Error Amplifier. It is internally connected to the inverting input of the PWM comparator. An optimal filter combination is chosen and connected to this pin and either ground or VFB to stabilize the voltage mode loop. UVLO input for VIN voltage. Connect a resistor divider between VIN and UVIN to set minimum operating voltage. Soft Start. Connect an external capacitor between SS and GND to set the soft start rate based on the 10A source current. The SS pin is held low via a 1mA (min) current during all fault conditions. Input connection to the high side N-channel MOSFET. Place a decoupling capacitor between this pin and PGND. Connect an inductor between this pin and VOUT No Connect High side driver supply pin. Connect BST to the external boost diode and capacitor as shown in the Typical Application Circuit on page 1. High side driver is connected between BST pin and SWN pin. Input for external 5V bias supply
5
VFB
6 7 9 10-13 14-16,23-26 17 18 22
COMP UVIN SS VIN LX NC BST Vcc
General Overview
THEORY OF OPERATION The SP7655 contains two unique control features that are very powerful in distributed applications. First, asynchronous driver control is enabled during startup, to prohibit the low side switch from pulling down the output until the high side switch has attempted to turn on. Second, a 100% duty cycle timeout ensures that the low side switch is periodically enhanced during extended periods at 100% duty cycle. This guarantees the synchronized refreshing of the BST capacitor during very large duty cycle ratios. The SP7655 also contains a number of valuable protection features. Programmable VIN UVLO allows the user to set the exact value at which the conversion voltage can safely begin down conversion, and an internal VCC UVLO which ensures that the controller itself has enough voltage to operate properly. Other protection fea(c) 2007 Sipex Corporation
The SP7655 is a fixed frequency, voltage mode, synchronous PWM regulator optimized for high efficiency. The part has been designed to be especially attractive for high voltage applications utilizing 5V to power the controller and 2.5V to 28V for step down conversion. The heart of the SP7655 is a wide bandwidth transconductance amplifier designed to accommodate Type II and Type III compensation schemes. A precision 0.8V reference, present on the positive terminal of the error amplifier, permits the programming of the output voltage down to 0.8V via the VFB pin. The output of the error amplifier, COMP, compared to a 1.1V peak-to-peak ramp is responsible for trailing edge PWM control. This voltage ramp, and PWM control logic are governed by the internal oscillator that accurately sets the PWM frequency to 300kHz.
Rev E: 3/2/07
SP7655 Wide Input Voltage Range 8A, 300kHz, Buck Regulator
THEORY OF OPERATION tures include thermal shutdown and short-circuit detection. In the event that either a thermal, short-circuit, or UVLO fault is detected, the SP7655 is forced into an idle state where the output drivers are held off for a finite period before a restart is attempted.
Soft Start Thermal and Short-Circuit Protection
Because the SP7655 is designed to drive large output current, there is a chance that the power converter will become too hot. Therefore, an internal thermal shutdown (145C) has been included to prevent the IC from malfunctioning at extreme temperatures. A short-circuit detection comparator has also been included in the SP7655 to protect against an accidental short at the output of the power converter. This comparator constantly monitors the positive and negative terminals of the error amplifier, and if the VFB pin falls more than 250mV (typical) below the positive reference, a short-circuit fault is set. Because the SS pin overrides the internal 0.8V reference during soft start, the SP7655 is capable of detecting shortcircuit faults throughout the duration of soft start as well as in regular operation.
Handling of Faults:
"Soft Start" is achieved when a power converter ramps up the output voltage while controlling the magnitude of the input supply source current. In a modern step down converter, ramping up the positive terminal of the error amplifier controls soft start. As a result, excess source current can be defined as the current required to charge the output capacitor. IVIN= C OUT * ( VOUT/ TSOFT-START)
The SP7655 provides the user with the option to program the soft start rate by tying a capacitor from the SS pin to GND. The selection of this capacitor is based on the 10A pullup current present at the SS pin and the 0.8V reference voltage. Therefore, the excess source can be redefined as: IVIN= C OUT * ( VOUT *10 A / (CSS * 0.8V)
Under Voltage Lock Out (UVLO)
The SP7655 contains two separate UVLO comparators to monitor the bias (VCC) and conversion (VIN) voltages independently. The VCC UVLO threshold is internally set to 4.25V, whereas the VIN UVLO threshold is programmable through the UVIN pin. When the UVIN pin is greater than 2.5V, the SP7655 is permitted to start up pending the removal of all other faults. Both the VCC and VIN UVLO comparators have been designed with hysteresis to prevent noise from resetting a fault.
Upon the detection of power (UVLO), thermal, or short-circuit faults, the SP7655 is forced into an idle state where the SS and COMP pins are pulled low and both switches are held off. In the event of UVLO fault, the SP7655 remains in this idle state until the UVLO fault is removed. Upon the detection of a thermal or short-circuit fault, an internal 200ms timer is activated. In the event of a short-circuit fault, a restart is attempted immediately after the 200ms timeout expires. Whereas, when a thermal fault is detected, the 200ms delay continuously recycles and a restart cannot be attempted until the thermal fault is removed and the timer expires.
Error Amplifier and Voltage Loop
Since the heart of the SP7655 voltage error loop is a high performance, wide bandwidth currentlimited (+/-150A) transconductance amplifier, there are many ways to compensate the voltage loop or to control the COMP pin externally. If a simple, single pole, single zero response is desired, then compensation can be
Rev E: 3/2/07
SP7655 Wide Input Voltage Range 8A, 300kHz, Buck Regulator
(c) 2007 Sipex Corporation
5
THEORY OF OPERATION as simple as an RC circuit to Ground. If a more complex compensation is required, then the amplifier has enough bandwidth (45 at 4 MHz) and enough gain (60dB) to run Type III compensation schemes with adequate gain and phase margins at crossover frequencies greater than 50kHz. The common mode output of the error amplifier is 0.9V to 2.2V. Therefore, the PWM voltage ramp has been set between 1.1V and 2.2V to ensure proper 0% to 100% duty cycle capability. The voltage loop also includes two other very important features. One is non-synchronous startup mode. Basically, the synchronous rectifier cannot turn on unless the high side switch has attempted to turn on or the SS pin has exceeded 1.7V. This feature prevents the controller from "dragging down" the output voltage during startup or in fault modes. The second feature is a 100% duty cycle timeout that ensures synchronized refreshing of the BST capacitor at very high duty ratios. In the event that the high side NFET is on for 20 continuous clock cycles, a reset is given to the PWM flipflop half way through the 21st cycle. This forces GL to rise for the cycle, in turn refreshing the BST capacitor. The boost capacitor is used to generate a high voltage drive supply for the high side switch, which is 5V above VIN.
Power MOSFETs
VBST GH Voltage VSWN V(VCC) GL Voltage 0V V(VIN) SWN Voltage -0V -V(Diode) V V(VIN)+V(VCC) BST Voltage V(VCC) TIME
Setting Output Voltages
The SP7655 can be set to different output voltages. The relationship in the following formula is based on a voltage divider from the output to the feedback pin VFB, which is set to an internal reference voltage of 0.80V. Standard 1% metal film resistors of surface mount size 0603 are recommended. Vout = 0.80V ( R1 / RSET + 1 ) => Rset = R1
.
The SP7655 contains a pair of integrated low resistance N-channel switches designed to drive up to 10 Amps of output current. Care should be taken to de-rate the output current based on the thermal conditions in the system such as ambient temperature, airflow and heat sinking. Maximum output current could be limited by thermal limitations of a particular application by taking advantage of the integrated-over-temperature protective scheme employed in the SP7655. The SP7655 incorporates a built-in over-temperature protection to prevent internal overheating.
Rev E: 3/2/07
[( Vout / 0.80V ) - 1 ]
Where R1 = 68.1K and for VOUT = 0.80V setting, simply remove RSET from the board. Furthermore, one could select the value of the R1 and RSET combination to meet the exact output voltage setting by restricting R1 resistance range such that 50K < R1 < 100K for overall system loop stability.
(c) 2007 Sipex Corporation
SP7655 Wide Input Voltage Range 8A, 300kHz, Buck Regulator
6
APPLICATIONS INFORMATION
Inductor Selection
There are many factors to consider in selecting the inductor including core material, inductance vs. frequency, current handling capability, efficiency, size and EMI. In a typical SP7655 circuit, the inductor is chosen primarily for value, saturation current and DC resistance. Increasing the inductor value will decrease output voltage ripple, but degrade transient response. Low inductor values provide the smallest size, but cause large ripple currents, poor efficiency and require more output capacitance to smooth out the larger ripple current. The inductor must be able to handle the peak current at the switching frequency without saturating, and the copper resistance in the winding should be kept as low as possible to minimize resistive power loss. A good compromise between size, loss and cost is to set the inductor ripple current to be within 20% to 40% of the maximum output current. The switching frequency and the inductor operating point determine the inductor value as follows:
I PEAK = I OUT (max) +
I PP 2
...and provide low core loss at the high switching frequency. Low cost powdered-iron cores have a gradual saturation characteristic but can introduce considerable AC core loss, especially when the inductor value is relatively low and the ripple current is high. Ferrite materials, although more expensive, have an abrupt saturation characteristic with the inductance dropping sharply when the peak design current is exceeded. Nevertheless, they are preferred at high switching frequencies because they present very low core loss while the designer is only required to prevent saturation. In general, ferrite or molypermalloy materials are a better choice for all but the most cost sensitive applications.
Optimizing Efficiency
L=
where:
VOUT (V IN (max) VOUT ) VIN (max) FS Kr I OUT ( max)
FS = switching frequency Kr = ratio of the AC inductor ripple current to the maximum output current The peak-to-peak inductor ripple current is:
The power dissipated in the inductor is equal to the sum of the core and copper losses. To minimize copper losses, the winding resistance needs to be minimized, but this usually comes at the expense of a larger inductor. Core losses have a more significant contribution at low output current where the copper losses are at a minimum, and can typically be neglected at higher output currents where the copper losses dominate. Core loss information is usually available from the magnetics vendor. Proper inductor selection can affect the resulting power supply efficiency by more than 15%! The copper loss in the inductor can be calculated using the following equation:
2 PL( Cu) = I L ( RMS ) RWINDING
I PP =
VOUT (VIN (max) VOUT ) VI N (max) FS L
Once the required inductor value is selected, the proper selection of core material is based on peak inductor current and efficiency requirements. The core must be large enough not to saturate at the peak inductor current...
Rev E: 3/2/07
where IL(RMS) is the RMS inductor current that can be calculated as follows: I PP IL(RMS)= I OUT(max) 1 + 1 3 IOUT(max)
()
2
SP7655 Wide Input Voltage Range 8A, 300kHz, Buck Regulator
(c) 2007 Sipex Corporation
7
APPLICATIONS INFORMATION
Output Capacitor Selection
The required ESR (Equivalent Series Resistance) and capacitance drive the selection of the type and quantity of the output capacitors. The ESR must be small enough that both the resistive voltage deviation due to a step change in the load current and the output ripple voltage do not exceed the tolerance limits expected on the output voltage. During an output load transient, the output capacitor must supply all the additional current demanded by the load until the SP7655 adjusts the inductor current to the new value. In order to maintain VOUT ,the capacitance must be large enough so that the output voltage is held up while the inductor current ramps to the value corresponding to the new load current. Additionally, the ESR in the output capacitor causes a step in the output voltage equal to the current. Because of the fast transient response and inherent 0% to100% duty cycle capability provided by the SP7655 when exposed to an output load transient, the output capacitor is typically chosen for ESR, not for capacitance value. The ESR of the output capacitor, combined with the inductor ripple current, is typically the main contributor to output voltage ripple. The maximum allowable ESR required to maintain a specified output voltage ripple can be calculated by: RESR where: VOUT = Peak-to-Peak Output Voltage Ripple IPK-PK= Peak-to-Peak Inductor Ripple Current The total output ripple is a combination of the ESR and the output capacitance value and can be calculated as follows: VOUT = IPP (1 - D) COUTFS
Rev E: 3/2/07
FS= Switching Frequency D = Duty Cycle COUT= Output Capacitance Value
Input Capacitor Selection
The input capacitor should be selected for ripple current rating, capacitance and voltage rating. The input capacitor must meet the ripple current requirement imposed by the switching current. In continuous conduction mode, the source current of the high-side MOSFET is approximately a square wave of duty cycle VOUT/VIN. More accurately the current wave form is trapezoidal, given a finite turn-on and turn-off, switch transition slope. Most of this current is supplied by the input bypass capacitors. The RMS current handling capability of the input capacitors is determined at maximum output current and under the assumption that the peak-to-peak inductor ripple current is low, it is given by: ICIN(RMS)= I OUT(max) D(1 - D) The worst case occurs when the duty cycle D is 50% and gives an RMS current value equal to IOUT/2. Select input capacitors with adequate ripple current rating to ensure reliable operation. The power dissipated in the input capacitor is:
2 P = ICIN ( rms ) R ESR( CIN ) CIN
IPK-PK
VOUT
This can become a significant part of power losses in a converter and hurt the overall energy transfer efficiency. The input voltage ripple primarily depends on the input capacitor ESR and capacitance. Ignoring the inductor ripple current, the input voltage ripple can be determined by:
VIN = I out (max) RE SR(CIN ) + I OUT ( MAX )VOUT (VI N VOUT ) FS C INV IN
2
(
)
2
+
(IPPRESR)2
SP7655 Wide Input Voltage Range 8A, 300kHz, Buck Regulator
(c) 2007 Sipex Corporation
8
APPLICATIONS INFORMATION The capacitor type suitable for the output capacitors can also be used for the input capacitors. However, exercise additional caution when tantalum capacitors are used. Tantalum capacitors are known for catastrophic failure when exposed to surge current, and input capacitors are prone to such surge current when power supplies are connected "live" to low impedance power sources. Although tantalum capacitors have been successfully employed at the input, it is generally not recommended.
Loop Compensation Design
The first step of compensation design is to pick the loop crossover frequency. High crossover frequency is desirable for fast transient response, but often jeopardizes the power supply stability. Crossover frequency should be higher than the ESR zero but less than 1/5 of the switching frequency or 60kHz. The ESR zero is contributed by the ESR associated with the output capacitors and can be determined by: Z(ESR)= 1 2 C OUT R ESR
The open loop gain of the whole system can be divided into the gain of the error amplifier, PWM modulator, buck converter output stage, and feedback resistor divider. In order to cross over at the desired frequency cut-off (FCO), the gain of the error amplifier compensates for the attenuation caused by the rest of the loop at this frequency. The goal of loop compensation is to manipulate loop frequency response such that its cross-over gain at 0db, results in a slope of 20db/dec.
Type III Voltage Loop Compensation GAMP (s) Gain Block VREF (Volts)
The next step is to calculate the complex conjugate poles contributed by the LC output filter, P(LC)= 1 2 L C OUT
When the output capacitors are of a Ceramic Type, the SP7655 Evaluation Board requires a Type III compensation circuit to give a phase boost of 180 in order to counteract the effects of an underdamped resonance of the output filter at the double pole frequency.
PWM Stage GPWMGain Block VIN VRAMP_PP Output Stage GOUT(s) Gain Block (SRESRCOUT+ 1) [S^2LCOUT+S(RESR+RDC) COUT+1]
+ _
(SRz2Cz2+1)(SR1Cz3+1) SR1Cz2(SRz3Cz3+1)(SRz2Cp1+1)
VOUT (Volts)
Notes: RESR= Output Capacitor Equivalent Series Resistance. RDC= Output Inductor DC Resistance. Condition: Cz2 >> Cp1 & R1 >> Rz3 Output Load Resistance >> RESR& R DC
VRAMP_PP= SP6132 Internal RAMP Amplitude Peak to Peak Voltage. Voltage Feedback GFBKGain Block R2 VFBK (Volts) (R1 + R2) or VREF
VOUT
SP7655 Voltage Mode Control Loop with Loop Dynamic
Definitions:
RESR= Output Capacitor Equivalent Series Resistance RDC= Output Inductor DC Resistance RRAMP_PP= SP7655 internal RAMP Amplitude Peak to Peak Voltage
Conditions: CZ2 >> Cp1 and R1 >> Rz3 Output Load Resistance >> RESRand RDC
Rev E: 3/2/07 SP7655 Wide Input Voltage Range 8A, 300kHz, Buck Regulator (c) 2007 Sipex Corporation
APPLICATIONS INFORMATION
SP765X Thermal Resistance
The SP765X family has been tested with a variety of footprint layouts along with different copper area and thermal resistance has been measured. The layouts were done on 4 layer FR4 PCB with the top and bottom layers using 3 ounces of copper and the power and ground layers using 1 ounce of copper. For the Minimum footprint, only about 0.1 square inch (of 3 ounces of) copper was used on the top or footprint layer, and this layer had no vias to connect to the 3 other layers. For the Medium footprint, about 0.7 square inches (of 3 ounces of) copper was used on the top layer, but vias were used to connect to the other 3 layers. For the Maximum footprint, about 1.0 square inch (of 3 ounces of) copper was used on the top layer and many vias were used to connect to the 3 other layers. The results show that only about 0.7 square inches (of 3 ounces of) copper on the top layer and vias connecting to the 3 other layers are needed to get the best thermal resistance of 36C/W. Adding area on the top beyond the 0.7 square inches did not reduce thermal resistance.
Using a minimum of 0.1 square inches of (3 ounces of) copper on the top layer with no vias connecting to the 3 other layers produced a thermal resistance of 44C/W. This thermal impedance is only 22% higher than the medium and large footprint layouts, indicating that space constrained designs can still benefit thermally from the Powerblox family of ICs. This indicates that a minimum footprint of 0.1 square inch, if used on a 4 layer board, can produce 44C/W thermal resistance. This approach is still very worthwhile if used in a space constrained design. The following page shows the footprint layouts from an ORCAD file. The thermal data was taken for still air, not with forced air. If forced air is used, some improvement in thermal resistance would be seen.
SP765X Thermal Resistance 4 Layer Board: Top Layer 3ounces Copper GND Layer 1ounce Copper Power Layer 1ounce Copper Bottom Layer 3ounces Copper Minimum Footprint: 44C/W Top Layer: 0.1 square inch No Vias to other 3 Layers Medium Footprint: 36C/W Top Layer: 0.7 square inch Vias to other 3 Layers Maximum Footprint: 36C/W Top Layer: 1.0 square inch Vias to other 3 Layers
Rev E: 3/2/07
SP7655 Wide Input Voltage Range 8A, 300kHz, Buck Regulator
(c) 2007 Sipex Corporation
0
Rev E: 3/2/07
SP7655 Wide Input Voltage Range 8A, 300kHz, Buck Regulator
(c) 2007 Sipex Corporation
APPLICATIONS INFORMATION
Gain (dB) Error Amplifier Gain Bandwidth Product
Condition: C22 >> CP1, R1 >> RZ3
20 Log (RZ2/R1)
1/6.28 (RZ2) (CP1) (RZ3) (CZ3)
1/6.28(R22) (CZ2)
1/6.28 (R1) (CZ3) (CZ2)
Frequency (Hz)
Bode Plot of Type III Error Amplifier Compensation.
CP1
RZ3 VOUT R1 68.1k, 1%
CZ3
CZ2 5 VFB
RZ2
+ + - 0.8V
R SET
6 COMP CF1
R SET
=54.48/ (VOUT -0.8) (k )
Type III Error Amplifier Compensation Circuit
Rev E: 3/2/07 SP7655 Wide Input Voltage Range 8A, 300kHz, Buck Regulator (c) 2007 Sipex Corporation
2
TYPICAL PERFORMANCE CHARACTERISTICS
100
Efficiency vs Load at 3.3Vin
100
Efficiency vs Load at 5.0Vin
90
90
80
Vout=2.5V Vout=1.8V
80
Vout=3.3V Vout=2.5V
70 Vout=1.2V
70
Vout=1.8V Vout=1.2V
60 0.0 1.0 2.0 3.0 4.0 5.0 6.0 7.0 8.0
60 0.0 1.0 2.0 3.0 4.0 5.0 6.0 7.0 8.0
Output Load (A)
Output Load (A)
100
Efficiency vs Load at 12Vin
100
Efficiency vs Load at 24Vin
90
90
80
Vout=5.0V Vout=3.3V
80
Vout=12V Vout=5.0V
70
Vout=2.5V Vout=1.8V Vout=1.2V
70
Vout=3.3V Vout=2.5V
60 0.0 1.0 2.0 3.0 4.0 5.0 6.0 7.0 8.0
60 0.00
1.00
2.00
3.00
4.00
5.00
6.00
7.00
8.00
Output Load (A)
Output Load (A)
Rev E: 3/2/07
SP7655 Wide Input Voltage Range 8A, 300kHz, Buck Regulator
(c) 2007 Sipex Corporation
3
TYPICAL PERFORMANCE CHARACTERISTICS
92.0 91.0
Efficiency (%)
SP7655 Effi. v.s Iout Plots @ Vin=12V, and Vout=3.3V
96.0 94.0
Efficiency (%)
SP7655 Effi. v.s Iout Plots @ Vin=12V, and Vout=5.0V
90.0 89.0 88.0 87.0 86.0 85.0 84.0 83.0 1 2 3 4 5 6 7 8 Load Current (A)
92.0 90.0 88.0 86.0 84.0 82.0 1 2 3 4 5 6 7 8
Load Current (A)
3.34
Output Voltage (V)
SP7655 Vout v.s Iout Plots @ Vin=12V, and Vout=3.3V
5.0000
Output Voltage (V)
SP7655 Vout v.s Iout Plots @ Vin=12V, and Vout=5.0V
3.33 3.32 3.31 3.3 0 1 2 3 4 5 6 7 8 Load Current (A)
4.9900 4.9800 4.9700 4.9600 0 1 2 3 4 5 6 7 8 Load Current (A)
Rev E: 3/2/07
SP7655 Wide Input Voltage Range 8A, 300kHz, Buck Regulator
(c) 2007 Sipex Corporation
TYPICAL PERFORMANCE CHARACTERISTICS
92.0 91.0 90.0 89.0 88.0 87.0 86.0 85.0 84.0 83.0 1 2
SP7655 Effi. v.s Iout Plots @ Vin=12V, and Vout=3.3V
96.0 94.0
SP7655 Effi. v.s Iout Plots @ Vin=12V, and Vout=5.0V
Efficiency (%)
Efficiency (%)
3 4 5 6 7 8
92.0 90.0 88.0 86.0 84.0 82.0 1 2 3 4 5 6 7 8
Load Current (A)
Load Current (A)
3.34 Output Voltage (V) 3.33 3.32 3.31 3.3 0 1
SP7655 Vout v.s Iout Plots @ Vin=12V, and Vout=3.3V
5.0000
Output Voltage (V)
SP7655 Vout v.s Iout Plots @ Vin=12V, and Vout=5.0V
4.9900 4.9800 4.9700 4.9600 0 1 2 3 4 5 6 7 8 Load Current (A)
Note: Fused Pin Area for pins -3 and pins -6 = (2e+b)xL - 2x(e-b) x L/2 = 0.376mm2 or 0.08 in2
2
3
4
5
6
7
8
Load Current (A)
Rev E: 3/2/07
SP7655 Wide Input Voltage Range 8A, 300kHz, Buck Regulator
(c) 2007 Sipex Corporation
5
Part Number
Temperature
ORDERING INFORMATION
Package
SP7655ER .............................................. -40C to +85C ................................. 26 Pin 7 X 4 DFN SP7655ER-L ........................................... -40C to +85C ............. (Lead Free) 26 Pin 7 X 4 DFN SP7655ER/TR ........................................ -40C to +85C ................................. 26 Pin 7 X 4 DFN SP7655ER-L/TR ..................................... -40C to +85C ............. (Lead Free) 26 Pin 7 X 4 DFN
Bulk Pack minimum quantity is 500. /TR = Tape and Reel. Pack quantity is 3,000 DFN.
Solved by
Sipex Corporation
TM
Headquarters and Sales Office 233 South Hillview Drive Milpitas, CA 5035 TEL: (08) 3-7500 FAX: (08) 35-7600
Sipex Corporation reserves the right to make changes to any products described herein. Sipex does not assume any liability arising out of the application or use of any product or circuit described herein; neither does it convey any license under its patent rights nor the rights of others.
Rev E: 3/2/07
SP7655 Wide Input Voltage Range 8A, 300kHz, Buck Regulator
(c) 2007 Sipex Corporation
6

▲Up To Search▲

Price & Availability of SP765507

	To Download SP765507 Datasheet File
If you can't view the Datasheet, Please click here to try to view without PDF Reader .